Communications system employing orthogonal chaotic spreading codes

ABSTRACT

Methods for code-division multiplex communications. The methods involve generating orthogonal or statistically orthogonal chaotic spreading codes (CSC 1 , . . . , CSC K ) using different sets of polynomial equations (f 0 (x(nT)), . . . , f N-1 (x(nT))), different constant values (C 0 , C 1 , . . . , C N-1 ) for the polynomial equations, or different sets of relatively prime numbers (p 0 , p 1 , . . . , p N-1 ) as modulus (m 0 , m 1 , . . . , m N-1 ) in solving the polynomial equations. The methods also involve forming spread spectrum communications signals using the orthogonal or statistically orthogonal chaotic spreading codes, respectively. The method further involve concurrently transmitting the spread spectrum communications signals over a common RF frequency band.

BACKGROUND OF THE INVENTION

1. Statement of the Technical Field

The invention concerns communications systems. More particularly, the invention concerns communications systems having a plurality of transmitters which communicate with corresponding receivers using spread spectrum waveforms.

2. Description of the Related Art

Pseudorandom number generators (PRNG) generally utilize digital logic or a digital computer and one or more algorithms to generate a sequence of numbers. While the output of conventional PRNG may approximate some of the properties of random numbers, they are not truly random. For example, the output of a PRNG has cyclo-stationary features that can be identified by analytical processes.

Chaotic systems can generally be thought of as systems which vary unpredictably unless all of its properties are known. When measured or observed, chaotic systems do not reveal any discernible regularity or order. Chaotic systems are distinguished by a sensitive dependence on a set of initial conditions and by having an evolution through time and space that appears to be quite random. However, despite its “random” appearance, chaos is a deterministic evolution.

Practically speaking, chaotic signals are extracted from chaotic systems and have random-like, non-periodic properties that are generated deterministically and are distinguishable from pseudo-random signals generated using conventional PRNG devices. In general, a chaotic sequence is one in which the sequence is empirically indistinguishable from true randomness absent some knowledge regarding the algorithm which is generating the chaos.

Some have proposed the use of multiple pseudo-random number generators to generate a digital chaotic-like sequence. However, such systems only produce more complex pseudo-random number sequences that possess all pseudo-random artifacts and no chaotic properties. While certain polynomials can generate chaotic behavior, it is commonly held that arithmetic required to generate chaotic number sequences requires an impractical implementation due to the precisions required.

Communications systems utilizing chaotic sequences offer promise for being the basis of a next generation of low probability of intercept (LPI) waveforms, low probability of detection (LPD) waveforms, and secure waveforms. While many such communications systems have been developed for generating chaotically modulated waveforms, such communications systems suffer from low throughput. The term “throughput”, as used herein, refers to the amount of data transmitted over a data link during a specific amount of time. This throughput limitation stems from the fact that a chaotic signal is produced by means of a chaotic analog circuit subject to drift.

The throughput limitation with chaos based communication systems can be traced to the way in which chaos generators have been implemented. Chaos generators have been conventionally constructed using analog chaotic circuits. The reason for reliance on analog circuits for this task has been the widely held conventional belief that efficient digital generation of chaos is impossible. Notwithstanding the apparent necessity of using analog type chaos generators, that approach has not been without problems. For example, analog chaos generator circuits are known to drift over time. The term “drift”, as used herein, refers to a slow long term variation in one or more parameters of a circuit. The problem with such analog circuits is that the inherent drift forces the requirement that state information must be constantly transferred over a communication channel to keep a transmitter and receiver synchronized.

The transmitter and receiver in coherent chaos based communication systems are synchronized by exchanging state information over a data link. Such a synchronization process offers diminishing return because state information must be exchanged more often between the transmitter and the receiver to obtain a high data rate. This high data rate results in a faster relative drift. In effect, state information must be exchanged at an increased rate between the transmitter and receiver to counteract the faster relative drift. Although some analog chaotic communications systems employ a relatively efficient synchronization process, these chaotic communications systems still suffer from low throughput.

The alternative to date has been to implement non-coherent chaotic waveforms. However, non-coherent waveform based communication systems suffer from reduced throughput and error rate performance. In this context, the phrase “non-coherent waveform” means that the receiver is not required to reproduce any synchronized copy of the chaotic signals that have been generated in the transmitter. The phrase “communications using a coherent waveform” means that the receiver is required to reproduce a synchronized copy of the chaotic signals that have been generated in the transmitter.

In view of the forgoing, there is a need for a coherent chaos-based communications system having an increased throughput. There is also a need for a chaos-based communications system configured for generating a signal having chaotic properties. As such, there is further a need for a chaos-based communications system that corrects drift between a transmitter and a receiver without an extreme compromise of throughput.

SUMMARY OF THE INVENTION

The present invention concerns communication systems and methods for code-division multiplex communications. Methods according to embodiments of the invention involve generating orthogonal or statistically orthogonal chaotic spreading codes using different sets of polynomial equations, respectively. The phrase “statistically orthogonal”, as used herein, means that the expectation value of an inner product of two random spreading sequences is zero. The sets of polynomial equations can differ with respect to at least one characteristic selected from the group comprising a constant and a degree. The methods also involve forming spread spectrum communications signals respectively using the orthogonal or statistically orthogonal chaotic spreading codes. The methods further involve concurrently transmitting the spread spectrum communication signals over a common RF frequency band. Subsequently, the spread spectrum communications signals are received at one or more receivers. At the receiver(s), chaotic de-spreading codes are generated and spread spectrum communication signals are de-spread using the chaotic de-spreading codes. Notably, the chaotic de-spreading codes are synchronized in time and frequency with the chaotic spreading codes, respectively.

According to an embodiment of the invention, the orthogonal chaotic spreading codes are generated using residue number system (RNS) arithmetic operations to respectively determine solutions for each set of polynomial equations. The solutions are iteratively computed and expressed as RNS residue values. The orthogonal chaotic spreading codes are generated by determining a series of digits in a weighted number system based on the RNS residue values. The orthogonal chaotic spreading codes are further generated by selecting a value for each of N moduli comprising a respective moduli set in an RNS used for respectively solving each set of polynomial equations. The moduli set can be different for each set of polynomial equations.

The code-division multiplex communications system include a plurality of transmitters. The transmitters are configured for generating orthogonal or statistically orthogonal chaotic spreading codes respectively using different sets of polynomial equations and/or different moduli sets. The sets of polynomial equations can differ with respect to at least one characteristic selected from the group comprising a constant and a degree. The transmitters are also configured for forming spread spectrum communications signals respectively using the orthogonal or statistically orthogonal chaotic spreading codes. The transmitters are further configured for concurrently transmitting the spread spectrum communication signals over a common RF frequency band.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments will be described with reference to the following drawing figures, in which like numerals represent like items throughout the figures, and in which:

FIG. 1 is a schematic illustration of a first exemplary coherent chaotic spread-spectrum communication system according to an embodiment of the invention.

FIG. 2 is a schematic illustration of a second exemplary coherent chaotic spread-spectrum communication system according to an embodiment of the invention.

FIG. 3 is a schematic illustration of a plurality of chaotic spreading codes according to embodiments of the invention.

FIG. 4 is a block diagram of the transmitter shown in FIGS. 1-2 according to an embodiment of the invention.

FIG. 5 is a block diagram of the radio frequency (RF) front end of FIGS. 1-2 according to an embodiment of the invention.

FIG. 6 is a block diagram of the receiver back end of FIGS. 1-2 according to an embodiment of the invention.

FIG. 7 is a block diagram of an exemplary radio frequency (RF) front end according to an embodiment of the invention.

FIG. 8 is a block diagram of an exemplary receiver back end according to an embodiment of the invention.

FIG. 9 is a conceptual diagram of the chaos generators of FIGS. 4, 6, and 8.

FIG. 10 is a flow diagram of a method for generating a chaotic spreading code (or chaotic sequence) according to an embodiment of the invention.

FIG. 11 is a block diagram of the chaos generator of FIG. 4 according to an embodiment of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Embodiments of the present invention will now be described with respect to FIGS. 1-11. Embodiments of the present invention relate to Code Division Multiplexing (CDM) based communications systems. CDM based communications systems according to embodiments of the present invention generally allow signals from a series of independent sources to be transmitted at the same time over the same frequency band. The signal transmissions are accomplished using orthogonal spreading codes to spread each signal over a large, common frequency band. The orthogonal spreading codes are advantageously distinct chaotic spreading codes generated by chaos generators employing different sets of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)), different sets of constants C₀, C₁, . . . , C_(N-1), and/or different sets of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected for use as modulus m₀, m₁, . . . , m_(N-1). The CDM based communications systems also allow transmitted signals to be received at one or more receivers. At the receivers, the appropriate orthogonal spreading codes are used to recover the original signals intended for a particular user.

It should be appreciated that the CDM based communications systems disclosed herein have many advantages as compared to conventional spread-spectrum communications systems. The CDM based communications systems disclosed herein also have many advantages over chaos based spread spectrum systems utilizing analog based chaotic sequence generators. For example, the CDM based communications systems provide output signals with a much smaller bandwidth as compared to the bandwidths of output signals generated by conventional code division multiple access (CDMA) based communications systems. This bandwidth efficiency results from the fact that the chaotic spreading signal has multilevel and random amplitudes. In one embodiment of the present invention, the amplitude distribution is Gaussian. As a result, the cross correlation between two spreading sequences is impulsive and the statistical orthogonality is more robust. The robust statistical orthogonality translates to a lower chipping rate for a fixed orthogonality versus conventional CDMA systems. In effect, there is a requirement for a reduced bandwidth. Conversely, the CDM based communications systems of the present invention can handle a relatively larger number of users in a fixed bandwidth. The CDM based communications systems disclosed herein also correct clock drifts between a transmitter and a receiver without an extreme compromise of throughput

Before describing the communications systems of the present invention, it will be helpful in understanding an exemplary environment in which the invention can be utilized. In this regard, it should be understood that the communications systems of the present invention can be utilized in a variety of different applications where the frequency re-use of a communications channel needs to be increased. Such applications include, but are not limited to, military applications and commercial mobile/cellular telephone applications.

Communications Systems

Referring now to FIG. 1, there is provided a schematic illustration of a first exemplary coherent chaotic spread-spectrum communication system 100 according to an embodiment of the invention. As shown in FIG. 1, the communication system 100 is comprised of a plurality of transmitters 102 ₁, . . . , 102 _(K) and a base station 104. The transmitters 102 ₁, . . . , 102 _(K) are generally configured to generate output signals having chaotic properties, i.e., output signals having their frequency spectrum varied over time. Each of the output signals are generated using a coherent chaotic sequence spread spectrum (CCSSS) method. The CCSSS method generally involves combining data symbols (e.g., phase shift keying symbols) with a higher rate chaotic spreading code CSC₁, . . . , CSC_(K). The chaotic spreading codes CSC₁, . . . , CSC_(K) are analogous to binary pseudo-noise spreading sequences or chipping codes employed by conventional direct sequence spread spectrum (DSSS) systems. The chaotic spreading codes CSC₁, . . . , CSC_(K) spread the spectrum of the data symbols according to a spreading ratio. The resulting signals resemble truly random signals.

Notably, the chaotic spreading codes CSC₁, . . . , CSC_(K) are distinct orthogonal or statistically orthogonal chaotic spreading codes. As noted above, the phrase “statistically orthogonal”, as used herein, means that the expectation value of an inner product of two random spreading sequences is zero. Methods for generating orthogonal or statistically orthogonal chaotic spreading codes will be described below in relation to FIGS. 9-11. However, it should be noted that orthogonal or statistically orthogonal chaotic spreading codes are generally generated by generators employing polynomial equations, constants, and/or relatively prime numbers as moduli. Accordingly, the distinct orthogonal or statistically orthogonal chaotic spreading codes are generally generated by chaos generators different sets of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)), different sets of constants C₀, C₁, . . . , C_(N-1), and/or different sets of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected for use as moduli m₀, m₁, . . . , m_(N-1).

A schematic illustration of exemplary distinct orthogonal or statistically orthogonal chaotic spreading codes CSC₁, . . . , CSC_(K) is provided in FIG. 3. As shown in FIG. 3, each of the orthogonal or statistically orthogonal chaotic spreading codes CSC₁, . . . , CSC_(K) is a different orthogonal or statistically orthogonal chaotic spreading code. For example, the orthogonal or statistically orthogonal chaotic spreading code CSC₁ is an orthogonal or statistically orthogonal chaotic spreading code [V₁ V₂ V₃ . . . V_(v-4) V_(v-3) V_(v-2) V_(v-1) V_(v)]. The orthogonal or statistically orthogonal chaotic spreading code CSC₂ is an orthogonal or statistically orthogonal chaotic spreading code [W₁ W₂ W₃ . . . W_(w-4) W_(w-3) W_(w-2) W_(w-1) W_(w)]. The orthogonal or statistically orthogonal chaotic spreading code CSC_(K) is an orthogonal or statistically orthogonal chaotic spreading code [Z₁ Z₂ Z₃ . . . Z_(z-4) Z_(z-3) Z_(z-2) Z_(z-1) Z_(z)]. The invention is not limited in this regard.

Referring again to FIG. 1, the series of independent transmitters 102 ₁, . . . , 102 _(K) are configured to transmit information (or output signals) to the base station 104. The information (or output signals) can be transmitted from the transmitters 102 ₁, . . . , 102 _(K) at the same time over the same communications channel 106 (or frequency band).

As shown in FIG. 1, the base station 104 is comprised of a radio frequency (RF) front end 108 ₁ and a plurality of receiver back ends 110 ₁, . . . , 110 _(K). The RF front end 108 is generally configured for receiving signals transmitted from the transmitters 102 ₁, . . . , 102 _(K), placing the received signals in a form suitable for processing by the receiver back ends 110 ₁, . . . , 110 _(K), and communicating the received signals to the receiver back ends 110 ₁, . . . , 110 _(K). Embodiments of the RF front end 108 will be described below in relation to FIG. 5 and FIG. 7.

The receiver back ends 110 ₁, . . . , 110 _(K) are configured for removing the randomness of the received signals to recover the original information (or data). In particular, the information (or data) is recovered by de-spreading the received signals using the appropriate orthogonal or statistically orthogonal chaotic spreading codes CSC₁, . . . , CSC_(K). In this regard, it should be understood that each of the receiver back ends 110 ₁, . . . , 110 _(K) is configured to generate a replica of a particular orthogonal or statistically orthogonal chaotic spreading code CSC₁, . . . , CSC_(K). For example, the receiver back end 110 ₁ is configured to generate a replica of the orthogonal or statistically orthogonal chaotic spreading code CSC₁ that is synchronized in time and frequency with the orthogonal or statistically orthogonal chaotic spreading code CSC₁. Similarly, the receiver back end 110 ₂ is configured to generate a replica of the orthogonal or statistically orthogonal chaotic spreading code CSC₂ that is synchronized in time and frequency with the orthogonal or statistically orthogonal chaotic spreading code CSC₂, and so on. Embodiments of the receiver back ends 110 ₁, . . . , 110 _(K) will be described below in relation to FIG. 6 and FIG. 8.

Referring now to FIG. 2, there is provided a schematic illustration of a second exemplary coherent chaotic spread-spectrum communication system 150 according to an embodiment of the invention. As shown in FIG. 2, the communication system 150 is comprised of a plurality of transmitters 102 ₁, . . . , 102 _(K) and a plurality of receivers 154 ₁, . . . , 154 _(K). The transmitters 102 ₁, . . . , 102 _(K) are the same as the transmitters of FIG. 1. As such, the description provided above in relation to the transmitters 102 ₁, . . . , 102 _(K) is sufficient for understanding the communication system 150.

Each of the receivers 154 ₁, . . . , 154 _(K) is comprised of an RF front end 108 ₁, . . . , 108 _(K) and a receiver back end 110 ₁, . . . , 110 _(K). The RF front ends 108 ₁, . . . , 108 _(K) are the same as or substantially similar to the RF front end 108 of FIG. 1. As such, the description provided above in relation to the RF front end 108 is sufficient for understanding the RF front ends 108 ₁, . . . , 108 _(K). Similarly, the receiver back ends 110 ₁, . . . , 110 _(K) are the same as the receiver back ends of FIG. 1. As such, the description provided above in relation to the receiver back ends 110 ₁, . . . , 110 _(K) is sufficient for understanding the communication system 150.

Transmitter Architectures

Referring now to FIG. 4, there is provided a block diagram of the transmitter 102 ₁ shown in FIGS. 1-2. The embodiment of the transmitter 102 ₁ assumes that: (1) a low order phase shift keying (PSK) data modulation is used; (2) no pulse shaping is applied to data symbols; (3) modulated data symbols are generated in quadrature form; and (4) chaotic spectral spreading is performed at a baseband intermediate frequency (IF). The transmitters 102 ₂, . . . , 102 _(K) are the same as or substantially similar to the transmitter 102 ₁. As such, the following description of the transmitter 102 ₁ is sufficient for understanding the transmitters 102 ₂, . . . , 102 _(K).

Referring again to FIG. 4, the transmitter 102 ₁ is generally configured for generating an amplitude-and-time-discrete baseband signal. The transmitter 102 ₁ is also configured for spreading the amplitude-and-time-discrete baseband signal over a wide intermediate frequency band. This spreading consists of multiplying the amplitude-and-time-discrete baseband signal by a digital chaotic sequence. The product of this arithmetic operation is hereinafter referred to as a digital chaotic signal. In this regard, it should be understood that the transmitter 102 ₁ is also configured to process the digital chaotic signal to place the same in a proper analog form suitable for transmission over a communications link. The transmitter 102 ₁ is further configured to communicate analog chaotic signals to a base station 104 (described above in relation to FIG. 1) and/or a receiver 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2) via a communications link.

As shown in FIG. 4, the transmitter 102 ₁ is comprised of a data source 402, a source encoder 404, a symbol formatter 406, an acquisition data generator 408, a transmitter controller 410, a multiplexer 414, a channel encoder 416, a precision real time reference 412, and a complex multiplier 424. The transmitter 102 ₁ is also comprised of a chaos generator 418, a real uniform statistics to quadrature Gaussian statistics mapper device (RUQG) 420, and a sample rate matching filter (SRMF) 422. The transmitter 102 is further comprised of an interpolator 426, a digital local oscillator (LO) 430, a real part of a complex multiplier 428, a digital-to-analog converter (DAC) 432, an anti-image filter 434, an intermediate frequency (IF) to radio frequency (RF) conversion device 436, and an antenna element 438.

The data source 402 is an interface configured for receiving an input signal containing data from an external device (not shown). As such, the data source 402 can be configured for receiving bits of data from the external data source (not shown). The data source 402 can further be configured for supplying bits of data to the source encoder 404 at a particular data transfer rate.

The source encoder 404 can be configured to encode the data received from the external device (not shown) using a forward error correction coding scheme. The bits of data received at or generated by the source encoder 404 represent any type of information that may be of interest to a user. For example, the data can be used to represent text, telemetry, audio, or video data. The source encoder 404 can further be configured to supply bits of data to the symbol formatter 406 at a particular data transfer rate.

The symbol formatter 406 is generally configured to process bits of data for forming channel encoded symbols. In a preferred embodiment, the source encoded symbols are phase shift keyed (PSK) encoded. If it is desired to use a non-coherent form of PSK with the coherent chaos spread spectrum system, then the symbol formatter 404 can also be configured for differentially encoding formed PSK symbols. Differential encoding is well known to persons having ordinary skill in the art, and therefore will not be described herein. The symbol formatter 406 can further be configured for communicating non-differentially encoded PSK symbols and/or differentially encoded PSK symbols to the multiplexer 414.

According to an embodiment of the invention, the symbol formatter 406 is functionally similar to a serial in/parallel out shift register where the number of parallel bits out is equal to log base two (log₂) of the order of the channel encoder 416. The symbol formatter 406 is selected for use with a quadrature phase shift keying (QPSK) modulator. As such, the symbol formatter 406 is configured for performing a QPSK formatting function for grouping two (2) bits of data together for a QPSK symbol (i.e., a single two bit parallel word). Thereafter, the symbol formatter 406 communicates the QPSK symbol data to the multiplexer 414. Still, the invention is not limited in this regard.

According to another embodiment of the invention, the symbol formatter 406 is functionally similar to a serial in/parallel out shift register where the number of parallel bits out is equal to log base two (log₂) of the order of the channel encoder 416. The symbol formatter 406 is selected for use with a binary phase shift keying (BPSK) modulator. As such, the symbol formatter 406 is configured for mapping one bit of data for a BPSK symbol. Thereafter, the symbol formatter 406 communicates the BPSK symbol data to the multiplexer 414. Still, the invention is not limited in this regard.

According to another embodiment of the invention, the symbol formatter 406 is selected for use with a sixteen quadrature amplitude modulation (16QAM) modulator. As such, the symbol formatter 406 is configured for mapping four (4) bits for a 16QAM symbol. Thereafter, the symbol formatter 406 communicates the 16QAM symbol data to the multiplexer 414. Still, the invention is not limited in this regard.

According to yet another embodiment of the invention, the symbol formatter 406 is selected for use with a binary amplitude shift keying (ASK) modulator. As such, the symbol formatter 406 is configured for mapping one bit of data for an ASK symbol. Thereafter, the symbol formatter 406 communicates the data for an ASK symbol to the multiplexer 414. Still, the invention is not limited in this regard.

Referring again to FIG. 4, the acquisition data generator 408 is configured for generating a “known data preamble”. The “known data preamble” can be a repetition of the same known symbol or a series of known symbols. The “known data preamble” can be used to enable initial synchronization of a chaotic sequence generated in the transmitter 102 ₁ and a base station 104 (described above in relation to FIG. 1) or receiver 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2). The duration of the “known data preamble” is determined by an amount required by a base station (described above in relation to FIG. 1) or receiver 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2) to synchronize with the transmitter 102 ₁ under known worst case channel conditions. The acquisition data generator 408 can be further configured for communicating the “known data preamble” to the multiplexer 414.

The multiplexer 414 is configured to receive a binary word (that is to be modulated by the channel encoder 416) from the symbol formatter 406. The multiplexer 414 is also configured to receive the “known data preamble” words from the acquisition data generator 408. The multiplexer 414 is coupled to the transmitter controller 410. The transmitter controller 410 is configured for controlling the multiplexer 414 so that the multiplexer 414 routes the “known data preamble” to the channel encoder 416 at the time of a new transmission.

According to alternative embodiments of the invention, the “known data preamble” is stored in a modulated form. In such a scenario, the architecture of FIG. 4 is modified such that the multiplexer 414 exists after the channel encoder 416. The “known data preamble” may also be injected at known intervals to aid in periodic resynchronization of the chaotic sequence generated in the transmitter 102 ₁ and a base station 104 (described above in relation to FIG. 1) or receiver 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2). This would typically be the case for an implementation meant to operate in harsh channel conditions. Still, the invention is not limited in this regard.

Referring again to FIG. 4, the multiplexer 414 can be configured for selecting data words to be routed to the channel encoder 416 after a preamble period has expired. The multiplexer 414 can also be configured for communicating data words to the channel encoder 416. In this regard, it should be appreciated that a communication of the data words to the channel encoder 416 is delayed by a time defined by the length of the “known data preamble.” This delay allows all of a “known data preamble” to be fully communicated to the channel encoder 416 prior to communication of the data words.

The channel encoder 416 can be configured for performing actions to represent the “known data preamble” and the data words in the form of a modulated amplitude-and-time-discrete digital signal symbols. The modulated amplitude-and-time-discrete digital signal symbols are defined by digital words which represent intermediate frequency (IF) modulated symbols comprised of bits of data having a one (1) value or a zero (0) value. Methods for representing digital symbols by an amplitude-and-time-discrete digital signal are well known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be appreciated that the channel encoder 416 can employ any known method for representing digital symbols by an amplitude-and-time-discrete digital signal.

As shown in FIG. 4, the channel encoder 416 can be selected as a digital baseband modulator employing quadrature phase shift keying (QPSK). As such, the output of the QPSK modulator includes an in-phase (“I”) data and quadrature phase (“Q”) data. Accordingly, the channel encoder 416 is configured for communicating I and Q data to the digital complex multiplier 424.

According an embodiment of the invention, the transmitter 102 ₁ is comprised of a sample rate matching device (not shown) between the channel encoder 416 and the complex multiplier 424. The sample rate matching device (not shown) can perform a sample rate increase on the amplitude-and-time-discrete digital signal so that a sample rate of the amplitude-and-time-discrete digital signal is the same as a digital chaotic sequence communicated to the digital complex multiplier 424. Still, the invention is not limited in this regard.

Referring again to FIG. 4, the digital complex multiplier 424 can be configured for performing a complex multiplication in the digital domain. In the digital complex multiplier 424, the amplitude-and-time-discrete digital signal from the channel encoder 416 is multiplied by a chaotic spreading code CSC₁. The chaotic spreading code CSC₁ is a digital representation of a chaotic sequence. The chaotic sequence is generated in the chaos generator 418. The chaos generator 418 is generally configured for generating the chaotic sequence in accordance with the methods described below in relation to FIGS. 9-10. Accordingly, the chaos generator 418 employs polynomial equations, constants, and/or relatively prime numbers as moduli for use in a chaotic sequence generation. The rate at which the digital chaotic sequence is generated can be an integer, rational or irrational multiple of a data symbol rate in one or more embodiments of the present invention. The greater the ratio between the data symbol period and the sample period of the digital chaotic sequence the higher a spreading gain. Notably, the chaos generator 418 can be configured for receiving initial conditions from the transmitter controller 410. The initial conditions define an arbitrary sequence starting location, i.e., the number of places (e.g., zero, one, two, Etc.) that a chaotic sequence is to be rotated to the right or cyclically shifted. The initial condition will be described below in relation to step 1014 of FIG. 10. The chaos generator 418 can also be configured for communicating the chaotic sequence to an RUQG 420.

The RUQG 420 can be configured for statistically transforming a digital chaotic sequence into a transformed digital chaotic sequence with pre-determined statistical properties. The transformed digital chaotic sequence can have a characteristic form including real, complex, and/or quadrature. The transformed digital chaotic sequence can have different word widths and/or different statistical distributions. For example, the RUQG 420 may take in two (2) uniformly distributed real inputs from the chaos generator 418 and convert those via a complex-valued bivariate Gaussian transformation to a quadrature output having statistical characteristics of a Guassian distribution. Such conversion techniques are well understood by those having ordinary skill in the art, and therefore will not be described in herein. However, it should be understood that such conversion techniques may use nonlinear processors, look-up tables, iterative processing (CORDIC functions), or other similar mathematical processes. The RUQG 420 can also be configured for communicating transformed chaotic sequences to the SRMF 422.

According to an embodiment of the invention, the RUQG 420 statistically transforms a digital chaotic sequence into a quadrature Gaussian form of the digital chaotic sequence. This statistical transformation is achieved via a nonlinear processor that combines lookup tables and embedded computational logic to implement the conversion of two (2) independent uniformly distributed random variables into a quadrature pair of Gaussian distributed variables. One such structure for this conversion is as shown in the mathematical equations (1) and (2).

G ₁=√{square root over (−2 log(u ₁))}·cos(2πu ₂)  (1)

G ₂=√{square root over (−2 log(u ₁))}·sin(2πu ₂)  (2)

where {u1, u2} are uniformly distributed independent input random variables and {G₁, G₂} are Gaussian distributed output random variables. The invention is not limited in this regard.

Referring again to FIG. 4, the SRMF 422 can be configured to resample the transformed chaotic sequence so that the chaos sample rate of the transformed chaotic sequence is at a preferred rate and matches a sample rate of the complex multiplier 424. The SRMF 422 can also be configured to communicate a resampled, transformed digital chaotic sequence to the digital complex multiplier 424.

According to an embodiment of the invention, the SRMF 422 comprises at least one real sample rate matching filter. The real sample rate matching filter is configured for resampling each of an in-phase processing path and a quadrature-phase processing path of the chaotic sequence. The real sample rate matching filter is also configured for communicating an in-phase (“I”) data and quadrature phase (“Q”) data to the digital complex multiplier 424. The invention is not limited in this regard.

Referring again to FIG. 4, the digital complex multiplier 424 is configured for performing complex-valued digital multiplication operations using the digital chaotic sequence output from the SRMF 422 and the amplitude-and-time-discrete digital signal output from the channel encoder 416. The result of the complex-valued digital multiplication operations is a digital representation of a coherent chaotic sequence spread spectrum modulated IF signal (hereinafter referred to as a “spread spectrum digital chaotic signal”). The spread spectrum digital chaotic signal comprises digital data that has been spread over a wide frequency bandwidth in accordance with a chaotic sequence generated by the chaos generator 418. The digital complex multiplier 424 is also configured to communicate spread spectrum digital chaotic signals to the interpolator 426.

The interpolator 426, real part of complex multiplier 428, and quadrature digital local oscillator 430 form at least one intermediate frequency (IF) translator. IF translators are well known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be understood that components 426, 428, 430 can be collectively configured for frequency modulating a digital chaotic signal received from the complex multiplier 424 to a sampled spread spectrum digital chaotic signal. The IF translator (i.e., component 428) is configured for communicating the sampled spread spectrum digital chaotic signal to the DAC 432, wherein the sampled spread spectrum digital chaotic signal has an increased sampling rate and a non-zero intermediate frequency. The DAC 432 can be configured for converting the sampled spread spectrum digital chaotic signal to an analog signal. The DAC 432 can also be configured for communicating the analog signal to the anti-image filter 434.

According to an embodiment of the invention, the complex multiplier 424 is configured for multiplying I and Q data of an amplitude-and-time-discrete digital signal by I and Q data of a digital chaotic sequence to obtain a spread spectrum digital chaotic signal. The spread spectrum digital chaotic signal is a quadrature, zero IF signal. The complex multiplier 424 is also configured for communicating the quadrature, zero IF signal to an IF translator. The IF translator comprises the interpolator 426, i.e., the IF translator is absent of the components 428, 430. The interpolator 426 is comprised of dual real interpolators configured for changing a sample rate of the quadrature, zero IF signal to a predetermined rate (e.g., seventy mega samples per second). The interpolator 426 communicates the sampled, quadrature, zero IF signal to the DAC 432. The DAC 432 is an interpolating DAC that increases the effective sample rate of the received signal (e.g., increases the predetermined rate to two hundred eighty mega samples per second). Interpolating DACs are well known to those having ordinary skill in the art, and therefore will not be described herein. The invention is not limited in this regard.

Referring again to FIG. 4, the anti-image filter 434 is configured for removing spectral images from the analog signal to form a smooth time domain signal. The anti-image filter 434 is also configured for communicating a smooth time domain signal to the RF conversion device 436. The RF conversion device 436 can be a wide bandwidth analog IF-to-RF up converter. The RF conversion device 436 is configured for forming an RF signal by centering a smooth time domain signal at an RF for transmission. The RF conversion device 436 is also configured for communicating RF signals to a power amplifier (not shown). The power amplifier (not shown) is configured for amplifying a received RF signal. The power amplifier (not shown) is also configured for communicating amplified RF signals to an antenna element 438 for communication to a base station 104 (described above in relation to FIG. 1) and/or a receiver 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2).

It should be understood that the digital generation of the digital chaotic sequence at the transmitter 102 ₁ and receiver (e.g., the base station 104 described above in relation to FIG. 1 or the receiver 154 ₁, . . . , 154 _(K) described above in relation to FIG. 2) is kept closely coordinated under the control of a precision real time reference 412 clock. If the precision of the clock 412 is relatively high, then the synchronization of the chaos generator 418 of the transmitter 102 ₁ and the chaos generator (described below in relation to FIG. 6 and FIG. 8) of the receiver (e.g., the base station 104 described above in relation to FIG. 1 or the receiver 154 ₁, . . . , 154 _(K) described above in relation to FIG. 2) is relatively close. The precision real time reference 412 allows the states of the chaos generators to be easily controlled with precision.

According to an embodiment of the invention, the precision real time reference 412 is a stable local oscillator locked to a precision real time reference (e.g., a global positioning system clock receiver or a chip scale atomic clock). The precision real time reference 412 is configured to supply a high frequency clock to the clocked logic circuits 404, . . . , 432 while being locked to a lower frequency reference clock. The lower frequency reference clock supplies a common reference and a common real time of day reference to prevent a large drift between the states of the chaos generator 418 and the chaos generator (described below in relation to FIG. 6 and FIG. 8) of the receiver (e.g., the base station 104 described above in relation to FIG. 1 or the receiver 154 ₁, . . . , 154 _(K) described above in relation to FIG. 2) over an extended time interval. The invention is not limited in this regard.

RF Front End and Receiver Back End Architectures

Referring now to FIG. 5, there is provided a more detailed block diagram of the RF front end 108 of FIG. 1. A more detailed block diagram of the receiver back end 110 ₁ of FIG. 1 is provided in FIG. 6. Notably, the RF front ends 108 ₁, . . . , 108 _(K) of FIG. 2 are the same as or substantially similar to the RF front end 108. As such the description provided below is sufficient for understanding the RF front ends 108 ₁, . . . , 108 _(K) of FIG. 12. Similarly, the receiver back ends 110 ₂, . . . , 110 _(K) of FIGS. 1-2 are the same as or substantially similar to the receiver back end 110 ₁. As such, the description provided below in relation to the receiver back end 110 ₁ is sufficient for understanding the receiver back ends 110 ₂, . . . , 110 _(K).

Referring now to FIG. 5, the RF front end 108 is generally configured for receiving transmitted analog chaotic signals from a transmitter 102 ₁, . . . , 102 _(K) (described above in relation to FIGS. 1-2 and FIG. 4). The RF front end 108 is also generally configured for down converting and digitizing a received analog chaotic signal. Accordingly, the RF front end 108 comprises an antenna element 502, a low noise amplifier (LNA) 504, a zonal filter 506, an automatic gain control (AGC) amplifier 508, a radio frequency (RF) to intermediate frequency (IF) conversion device 510, an anti-alias filter 512, and an analog-to-digital (A/D) converter 514.

Antenna element 502 is generally configured for receiving an analog input signal communicated from transmitter 102 ₁ over a communications link. Antenna element 502 can also be configured for communicating the analog input signal to LNA 504. LNA 504 is generally configured for amplifying a received analog input signal while adding as little noise and distortion as possible. LNA 504 can also be configured for communicating an amplified, analog input signal to zonal filer 506. Zonal filter 506 is configured for suppressing large interfering signals outside of bands of interest. Zonal filter 506 can also be configured for communicating filtered, analog input signals to the AGC amplifier 508. AGC amplifier 508 is generally a controllable gain amplifier configured for adjusting a gain of an analog input signal. AGC amplifier 508 is configured for communicating gain adjusted, analog input signals to the RF-to-IF conversion device 510.

The RF-to-IF conversion device 510 is generally configured for mixing an analog input signal to a particular IF. The RF-to-IF conversion device 510 is also configured for communicating mixed analog input signals to the anti-alias filter 512. Anti-alias filter 512 is configured for restricting a bandwidth of a mixed analog input signal. Anti-alias filter 512 is also configured for communicating filtered, analog input signals to the A/D converter 514. A/D converter 514 is configured for converting received analog input signals to digital signals. A/D converter 514 is also configured for communicating digital input signals to one or more receiver back ends (e.g., the receiver back ends 110 ₁, . . . , 110 _(K)).

Referring now to FIG. 6, the receiver back end 110 ₁ is generally configured for de-spreading a transmitted analog chaotic signal by correlating it with a replica of the chaotic sequence generated at a transmitter 102 ₁, . . . , 102 _(K). Notably, the replica chaotic sequence is time synchronized to the transmitted analog chaotic signal, i.e., a sampling rate of the replica chaotic sequence is the same as a sampling rate of the transmitted analog chaotic signal and is synchronized with a clock (not shown) of the transmitter 102 ₁, . . . , 102 _(K). The receiver back end 110 ₁ is further configured for processing de-spreaded analog chaotic signals to obtain data contained therein. The data can be converted into text, sound, pictures, navigational-position information, and/or any other type of useful payload information that can be communicated.

Notably, the receiver back end 110 ₁ of FIG. 6 is designed to eliminate the drawbacks of conventional analog based coherent communications systems. In this regard, it should be understood that analog chaos circuits of conventional analog based coherent communications systems are synchronized by periodically exchanging state information. The exchange of state information requires a substantial amount of additional bandwidth. In contrast, the receiver back end 110 ₁ is configured to synchronize two (2) strings of discrete time chaotic samples (i.e., chaotic sequences) without using a constant or periodic transfer of state update information. This synchronization feature of the receiver back end 110 ₁ will become more apparent as the discussion progresses.

As shown in FIG. 6, the receiver back end 110 ₁ comprises a Quadrature Fixed Digital Local Oscillator (QFDLO) 608, real multipliers 610, 612, Low Pass Filters (LPFs) 614, 616, a complex multiplier 618, a loop control circuit 620, a quadrature digital local oscillator 622, a correlator 628, a multiplexers 646, 648, a Channel Encoded Acquisition Data Generator (CEADG) 650, complex multipliers 624, 652, and a symbol timing recovery circuit 626. The receiver back end 110 ₁ also comprises a receiver controller 638, a precision real time reference clock 636, a hard decision device 630, a symbol to bits (S/B) converter 632, and a source decoder 634. The receiver back end 110 ₁ further comprises a chaos generator 640, a Real Uniform statistic to Quadrature Gaussian statistic mapper (RUQG) 642, and a re-sampling filter 644.

The QFDLO 608, real multipliers 610, 612 and LPFs 614, 616 combine to form a digital Weaver modulator. The digital Weaver modulator forms a baseband quadrature signal from the real IF signal generated by the RF front end 108. The quadrature digital local oscillator 622 is generally configured for generating a complex quadrature amplitude-and-time-discrete digital sinusoid at selectable phases and frequencies to fine tune the baseband quadrature signal. The digital sinusoid can be generated using a binary phase control word and a binary frequency control word received from the loop control circuit 620. The quadrature digital local oscillator 622 is also configured for communicating digital words representing quadrature digital sinusoid to the complex multiplier 618.

The complex multiplier 618 is configured for receiving digital words from the LPFs 614, 616 and digital words from the quadrature digital local oscillator 622. The complex multiplier 618 is also configured for generating digital output words by multiplying digital words from the LPFs 614, 616 by digital words from the quadrature digital local oscillator 622. The complex multiplier 618 is further configured for communicating data represented as digital output words to the complex multiplier 624 and the correlator 628.

The complex multiplier 624 is configured for performing a complex multiplication in the digital domain. The complex multiplication can involve multiplying digital words received from the complex multiplier 618 by digital words representing a chaotic sequence. The chaotic sequence is generated in the chaos generator 640. The chaos generator 640 is generally configured for generating the chaotic sequence in accordance with the methods described below in relation to FIGS. 9-10. Accordingly, the chaos generator 640 employs polynomial equations, constants, and/or relatively prime numbers as moduli for use in a chaotic sequence generation.

The chaos generator 640 is also configured for communicating chaotic sequences to the RUQG 642. In this regard, it should be appreciated that the chaos generator 640 is coupled to the receiver controller 638. The receiver controller 638 is configured to control the chaos generator 640 so that the chaos generator 640 generates a chaotic sequence with the correct initial state when the receiver back end 110 ₁ is in an acquisition mode and corrected states in a tracking mode.

The RUQG 642 can be configured for statistically transforming digital chaotic sequences into transformed digital chaotic sequences. Each of the transformed digital chaotic sequences can have a characteristic form. The characteristic form can include, but is not limited to, real, complex, quadrature, and combinations thereof. Each of the transformed digital chaotic sequences can have different word widths and/or different statistical distributions. The RUQG 642 is also configured for communicating transformed chaotic sequences to the re-sampling filter 644.

According to the embodiment of the present invention, the RUQG 642 is configured for statistically transforming a real uniform digital chaotic sequence into a quadrature Gaussian form of the digital chaotic sequence. The RUQG 642 is also configured for communicating the quadrature Gaussian form of the digital chaotic sequence to the re-sampling filter 644. More particularly, the RUQG 642 communicates an in-phase (“I”) data and quadrature phase (“Q”) data to the re-sampling filter 644. Embodiments of the present invention are not limited in this regard.

Referring again to FIG. 6, the re-sampling filter 644 is configured for forwarding transformed chaotic sequences to the digital complex multiplier 624. The re-sampling filter 644 is also configured for making a chaos sample rate compatible with a received signal sample rate when the receiver back end 110 ₁ is in acquisition mode. The re-sampling filter 644 is further configured to compensate for transmit and receive clock offsets with less than a certain level of distortion when the receiver back end 110 ₁ is in a steady state demodulation mode. In this regard, it should be appreciated that the re-sampling filter 644 is configured for converting a sampling rate of in-phase (“I”) and quadrature-phase (“Q”) data sequences from a first sampling rate to a second sampling rate without changing the spectrum of the data contained therein. The re-sampling filter 644 is configured to communicate in-phase (“I”) and quadrature-phase (“Q”) data sequences to the digital complex multipliers 624, 652 and the multiplexers 646, 648.

It should be noted that if a sampled form of a chaotic sequence is thought of as discrete samples of a continuous band limited chaos then the re-sampling filter 644 is effectively tracking the discrete time samples, computing a continuous representation of the chaotic sequence, and re-sampling the chaotic sequence at the discrete time points required to match the discrete time points sampled by the A/D converter 514. In effect, input values and output values of the re-sampling filter 644 are not exactly the same because the values are samples of the same waveform taken at slightly offset times. However, the values are samples of the same waveform so the values have the same power spectral density.

Referring again to FIG. 6, CEADG 650 is configured for generating modulated acquisition sequences. CEADG 650 is also configured for communicating modulated acquisition sequences to the complex multiplier 652. The complex multiplier 652 is configured for performing complex multiplication in the digital domain to yield a reference for a digital input signal. This complex multiplication can involve multiplying a modulated acquisition sequence received from the CEADG 650 by a digital representation of a chaotic sequence. The digital complex multiplier 652 is also configured for communicating reference signals to the multiplexers 646, 648.

Multiplexer 646 is configured for routing the quadrature-phase part of a reference signal to the correlator 628. Similarly, the multiplexer 648 is configured for routing the in-phase part of a reference signal to the correlator 628. In this regard, it should be appreciated that the multiplexers 646, 648 are coupled to the receiver controller 638. The receiver controller 638 is configured for controlling the multiplexers 646, 648 in tandem so that the multiplexers 646, 648 route the reference signal to the correlator 628 while the receiver back end 110 ₁ is in an acquisition mode (described below).

Correlator 628 is configured for correlating a chaotic sequence with a digital input signal. In this regard, it should be understood that, the sense of the real and imaginary components of the correlation is directly related to the values of the real and imaginary components of the symbols of a digital input signal. It should also be understood that the sense of the real and imaginary components of the correlation can be directly related to the values of the real and imaginary components of the PSK symbols of a digital input signal. Thus when the correlator 628 is in a steady state demodulation mode, the output of the correlator 628 is PSK symbol soft decisions. The phrase “soft decisions”, as used herein, refers to soft-values (which are represented by soft-decision bits) that comprise information about the bits contained in a sequence. Soft-values are values that represent the probability that a particular bit in a sequence is either a one (1) or a zero (0). For example, a soft-value for a particular bit can indicate that a probability of a bit being a one (1) is p(1)=0.3. Conversely, the same bit can have a probability of being a zero (0) which is p(0)=0.7.

Correlator 628 is also configured for communicating PSK soft decisions to the hard decision device 630 or source decoder 634 for final symbol decision making. The hard decision device 630 is configured for communicating symbol decisions to the S/B converter 632. S/B converter 632 is configured for converting symbols to a binary form. S/B converter 632 is also configured for communicating a binary data sequence to the source decoder 634. Source decoder 634 is configured for decoding FEC applied at a transmitter (e.g. the transmitter 102 ₁ described above in relation to FIGS. 1-2 and FIG. 4). Source decoder 634 is also configured for passing decoded bit streams to one or more external devices (not shown) utilizing the decoded data.

Correlator 628 is generally configured for acquiring initial timing information associated with a chaotic sequence and initial timing associated with a data sequence. Correlator 628 is further configured for tracking phase and frequency offset information between a chaotic sequence and a digital input signal and for tracking input signal magnitude information between the chaotic sequence and the digital input signal. Methods for acquiring initial timing information are well known to persons having ordinary skill in the art, and therefore will not be described herein. Similarly, methods for tracking phase/frequency offset information are well known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be appreciated that any such method for acquiring initial timing information and/or for tracking phase/frequency offset information can be used without limitation.

Correlator 628 is configured for communicating magnitude and phase information as a function of time to the loop control circuit 620. Loop control circuit 620 is configured for using magnitude and phase information to calculate a deviation of an input signal magnitude from a nominal range and to calculate phase/frequency offset information. The calculated information can be used to synchronize a chaotic sequence with a digital input signal. Loop control circuit 620 is also configured for communicating phase/frequency offset information to the quadrature digital local oscillator 622 and for communicating gain deviation compensation information to the AGC amplifier 608. Loop control circuit 620 is further configured for communicating retiming control signals to the re-sampling filter 644 and the chaos generator 640.

Precision real time reference 636 is the same as or substantially similar to the precision real time reference 412 of FIG. 4. The description provided above in relation to the precision real time reference 412 is sufficient for understanding the precision real time reference 636 of FIG. 6.

The operation of the receiver back end 110 ₁ will now be briefly described with regard to an acquisition mode and a steady state demodulation mode.

Acquisition Mode:

In acquisition mode, the re-sampling filter 644 performs a rational rate change and forwards a transformed chaotic sequence to the digital complex multiplier 652. The CEADG 650 generates a modulated acquisition sequence and forwards the same to the digital complex multiplier 652. The digital complex multiplier 652 performs a complex multiplication in the digital domain. In the digital complex multiplier 652, a modulated acquisition sequence from the CEADG 650 is multiplied by a digital representation of a chaotic sequence to yield a reference for a digital input signal that was generated at a transmitter (e.g., the transmitter 102 ₁ described above in relation to FIGS. 1-2 and FIG. 4) to facilitate initial acquisition. The chaotic sequence is generated in the chaos generator 640. The digital complex multiplier 652 communicates a reference signal to the multiplexers 646, 648. The multiplexers 646, 648 route the reference signal to the correlator 628. The correlator 628 is transitioned into a search mode. In this search mode, the correlator 628 searches across an uncertainty window to locate a received signal state so that the chaos generator 640 can be set with the time synchronized state vector.

Steady State Demodulation Mode:

In steady state demodulation mode, the correlator 628 tracks the correlation between the received modulated signal and the locally generated chaos close to the nominal correlation peak to generate magnitude and phase information as a function of time. This information is passed to the loop control circuit 620. The loop control circuit 620 applies appropriate algorithmic processing to this information to extract phase offset, frequency offset, and magnitude compensation information. The correlator 628 also passes its output information, based on correlation times terminated by symbol boundaries, to the hard decision block 630.

The hard decision block 630 compares the correlation information to pre-determined thresholds to make hard symbol decisions. The loop control circuit 620 monitors the output of the correlator 628. When the loop control circuit 620 detects fixed correlation phase offsets, the phase control of the quadrature digital local oscillator 622 is modified to remove the phase offset. When the loop control circuit 620 detects phase offsets that change as a function of time, it adjusts the re-sampling filter 644 which acts as an incommensurate re-sampler when the receiver back end 110 ₁ is in steady state demodulation mode or the frequency control of the quadrature digital local oscillator 622 is modified to remove frequency or timing offsets.

When the correlator's 628 output indicates that the received digital input signal timing has “drifted” more than plus or minus a half (½) of a sample time relative to a locally generated chaotic sequence, the loop control circuit 620 (1) adjusts a correlation window in an appropriate temporal direction by one sample time, (2) advances or retards a state of the local chaos generator 640 by one iteration state, and (3) adjusts the re-sampling filter 644 to compensate for the time discontinuity. This loop control circuit 620 process keeps the chaos generator 418 of the transmitter (e.g., transmitter 102 ₁ described above in relation to FIG. 4) and the chaos generator 640 of the receiver back end 110 ₁ synchronized to within half (½) of a sample time.

If a more precise temporal synchronization is required to enhance performance, a re-sampling filter can be implemented as a member of the class of polyphase fractional time delay filters. This class of filters is well known to persons having ordinary skill in the art, and therefore will not be described herein.

As described above, a number of chaotic samples are combined with an information symbol at the transmitter (e.g., the transmitter 102 ₁). Since the transmitter (e.g., the transmitter 102 ₁) and receiver back end 110 ₁ timing are referenced to two (2) different precision real time reference clock 412, 636 oscillators, symbol timing must be recovered at the receiver back end 110 ₁ to facilitate robust demodulation. Symbol timing recovery can include (1) multiplying a received input signal by a complex conjugate of a locally generated chaotic sequence using the complex multiplier 624, (2) computing an N point running average of the product where N is a number of chaotic samples per symbol time, (3) storing the values, the maximum absolute values of the running averages, and the time of occurrence, and (4) statistically combining the values at the symbol timing recovery circuit 626 to recover symbol timing. It should be noted that symbol timing recovery can also be accomplished via an output of the correlator 628. However, additional correlator operations are needed in such a scenario. As should be appreciated, using a separate multiplier operation for this purpose adds additional capabilities to the receiver (e.g., the base station 104 of FIG. 1 and the receivers 154 ₁, . . . , 154 _(K) of FIG. 2). The additional capabilities include, but are not limited to, the capability to correlate and post process over multiple correlation windows simultaneously to locate the best statistical fit for symbol timing.

In this steady state demodulation mode, the symbol timing recovery circuit 626 communicates a symbol onset timing to the correlator 628 for controlling an initiation of a symbol correlation. Correlator 628 correlates a locally generated chaotic sequence with a received digital input signal during a symbol duration. The sense and magnitude of a real and imaginary components of the correlation are directly related to the values of the real and imaginary components of symbols of a digital input signal. Accordingly, correlator 628 generates symbol soft decisions. Correlator 628 communicates the symbol soft decisions to the hard decision device 630 for final symbol decision making. Hard decision device 630 determines symbols using the symbol soft decisions. Thereafter, hard decision device 630 communicates the symbols to the S/B converter 632. S/B converter 632 converts the symbol decisions to a binary form. S/B converter 632 communicates a binary data sequence to the source decoder 634. Source decoder 634 decides FEC applied at the transmitter (e.g., the transmitter 102 ₁ described above in relation to FIGS. 1-2 and FIG. 4). Source decoder 634 also passes the decoded bit stream to one or more external devices (not shown) utilizing the decoded data.

Referring now to FIG. 7, there is provided a block diagram of another exemplary embodiment of an RF front end 700. Another exemplarily embodiment of a receiver back end is provided in FIG. 8. As shown in FIG. 7, the RF front end 700 is comprised of a plurality of components 702, 704, 706, 708, 710, 712. The components 702, 704, 706, 710, 712 of the RF front end 700 of FIG. 7 are the same as or substantially similar to the respective components 502, 504, 506, 512, 514 of FIG. 5. As such, the description provided above in relation to the components 502, 504, 506, 512, 514 is sufficient for understanding the components 702, 704, 706, 710, 712 of the RF front end 700. Component 708 of the RF front end 700 is an IF translator. IF translators are well known to those having ordinary skill in the art, and therefore will not be described herein.

As shown in FIG. 8, the receiver back end 800 is comprised of a loop control circuit 866, a correlator 868, and a digital complex multiplier 870. The receiver back end 800 is also comprised of a receiver controller 874, a precision real time reference 876, a hard decision device 872, a symbol to bits (S/B) converter 884, and a source decoder 886. The receiver back end 800 is further comprised of a residue number system (RNS) chaos generator 882 and a real uniform statistics to quadrature Gaussian statistics mapper 878. Each of the above listed components 854-886, 892 are similar to the respective components 602-606, 612, 614, 620, 628-642, 652 of FIG. 6. Thus, the description provided above in relation to components 602-506, 612, 614, 620, 628-642, 652 is sufficient for understanding the components 854-886, 892 of the receiver back end 800.

Chaos Generators and Digital Chaotic Sequence Generation

Referring now to FIG. 9, there is provided a conceptual diagram of a chaos generator 418, 640, 882 (described above in relation to FIG. 4, FIG. 6, and FIG. 8). As shown in FIG. 9, generation of the chaotic sequence begins with N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)). The polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be selected as the same polynomial equation or as different polynomial equations. According to an aspect of the invention, the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) are selected as irreducible polynomial equations having chaotic properties in Galois field arithmetic. Such irreducible polynomial equations include, but are not limited to, irreducible cubic polynomial equations and irreducible quadratic polynomial equations. The phrase “irreducible polynomial equation”, as used herein, refers to a polynomial equation that cannot be expressed as a product of at least two nontrivial polynomial equations over the same Galois field (f). For example, the polynomial equation f(x(nT)) is irreducible if there does not exist two (2) non-constant polynomial equations g(x(nT)) and h(x(nT)) in x(nT) with rational coefficients such that f(x(nT))=g(x(nT))·h(x(nT)).

Each of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be solved independently to obtain a respective solution. Each solution can be expressed as a residue number system (RNS) residue value using RNS arithmetic operations, i.e., modulo operations. Modulo operations are well known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be appreciated that an RNS residue representation for some weighted value “a” can be defined by mathematical equation (3).

R={a modulo m₀, a modulo m₁, . . . , a modulo m_(N-1)}  (3)

where R is a RNS residue N-tuple value representing a weighted value “a” and m₀, m₁, . . . , m_(N-1) respectively are the moduli for RNS arithmetic operations applicable to each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). R(nT) can be a representation of the RNS solution of a polynomial equation f(x(nT)) defined as R(nT)={f₀(x(nT)) modulo m₀, f₁(x(nT)) modulo m₁, . . . , f_(N-1)(x(nT)) modulo m_(N-1)}.

From the foregoing, it will be appreciated that the RNS employed for solving each of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) respectively has a selected modulus value m₀, m₁, . . . , m_(N-1). The modulus value chosen for each RNS moduli is preferably selected to be relatively prime numbers p₀, p₁, . . . , p_(N-1). The phrase “relatively prime numbers”, as used herein, refers to a collection of natural numbers having no common divisors except one (1). Consequently, each RNS arithmetic operation employed for expressing a solution as an RNS residue value uses a different relatively prime number p₀, p₁, . . . , p_(N-1) as a moduli m₀, m₁, . . . , m_(N-1).

The RNS residue value calculated as a solution to each one of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) will vary depending on the choice of prime numbers p₀, p₁, . . . , p_(N-1) selected as a moduli m₀, m₁, . . . , m_(N-1). Moreover, the range of values will depend on the choice of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected as a moduli m₀, m₁, . . . , m_(N-1). For example, if the prime number five hundred three (503) is selected as modulus m₀, then an RNS solution for a first polynomial equation f₀(x(nT)) will have an integer value between zero (0) and five hundred two (502). Similarly, if the prime number four hundred ninety-one (491) is selected as modulus m₁, then the RNS solution for a second polynomial equation f₁(x(nT)) has an integer value between zero (0) and four hundred ninety (490).

According to an embodiment of the invention, each of the polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) is selected as an irreducible cubic polynomial equation having chaotic properties in Galois field arithmetic. Each of the N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can also be selected to be a constant or varying function of time. The irreducible cubic polynomial equation is defined by a mathematical equation (4).

f(x(nT))=Q(k)x ³(nT)+R(k)x ²(nT)+S(k)x(nT)+C(k,L)  (4)

where: x is a variable defining a sequence location; n is a sample time index value; k is a polynomial time index value; L is a constant component time index value; T is a fixed constant having a value representing a time interval or increment; Q, R, and S are coefficients that define the polynomial equation f(x(nT)); and C is a coefficient of x(nT) raised to a zero power and is therefore a constant for each polynomial characteristic.

In a preferred embodiment, a value of C is selected which empirically is determined to produce an irreducible form of the stated polynomial equation f(x(nT)) for a particular prime modulus. For a given polynomial with fixed values for Q, R, and S more than one value of C can exist, each providing a unique iterative sequence. Still, the invention is not limited in this regard.

According to another embodiment of the invention, the N polynomial equations f₀(x(nT)) . . . f_(N-1)(x(nT)) are identical exclusive of a constant value C. For example, a first polynomial equation f₀(x(nT)) is selected as f₀(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C₀. A second polynomial equation f₁(x(nT)) is selected as f₁(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C₁. A third polynomial equation f₂(x(nT)) is selected as f₂(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C₂, and so on. Each of the constant values C₀, C₁, . . . , C_(N-1) is selected to produce an irreducible form in a residue ring of the stated polynomial equation f(x(nT))=3x³(nT)+3x²(nT)+x(nT)+C. In this regard, it should be appreciated that each of the constant values C₀, C₁, . . . , C_(N-1) is associated with a particular modulus m₀, m₁, . . . , m_(N-1) value to be used for RNS arithmetic operations when solving the polynomial equation f(x(nT)). Such constant values C₀, C₁, . . . , C_(N-1) and associated modulus m₀, m₁, . . . , m_(N-1) values which produce an irreducible form of the stated polynomial equation f(x(nT)) are listed in the following Table (1).

TABLE 1 Moduli values Sets of constant values m₀, m₁, . . . , m_(N−1): C₀, C₁, . . . , C_(N−1): 3 {1, 2} 5 {1, 3} 11 {4, 9} 29 {16, 19} 47 {26, 31} 59 {18, 34} 71 {10, 19, 20, 29} 83 {22, 26, 75, 79} 101 {27, 38, 85, 96} 131 {26, 39, 77, 90} 137 {50, 117} 149 {17, 115, 136, 145} 167 {16, 32, 116, 132} 173 {72, 139} 197 {13, 96, 127, 179} 233 {52, 77} 251 {39, 100, 147, 243} 257 {110, 118} 269 {69, 80} 281 {95, 248} 293 {37, 223} 311 {107, 169} 317 {15, 55} 347 {89, 219} 443 {135, 247, 294, 406} 461 {240, 323} 467 {15, 244, 301, 425} 479 {233, 352} 491 {202, 234} 503 {8, 271} Still, embodiments of the present invention are not limited in this regard.

The number of discrete magnitude states (dynamic range) that can be generated with the system shown in FIG. 9 will depend on the quantity of polynomial equations N and the modulus values m₀, m₁, . . . , m_(N-1) values selected for the RNS number systems. In particular, this value can be calculated as the product M=m₀·m₁,·m₃·m₄· . . . ·m_(N-1).

Referring again to FIG. 9, it should be appreciated that each of the RNS solutions No. 1, . . . , No. N is expressed in a binary number system representation. As such, each of the RNS solutions No. 1, . . . , No. N is a binary sequence of bits. Each bit of the sequence has a zero (0) value or a one (1) value. Each binary sequence has a bit length selected in accordance with a particular moduli.

According to an embodiment of the invention, each binary sequence representing a residue value has a bit length (BL) defined by a mathematical equation (5).

BL=Ceiling[Log 2(m)]  (5)

where m is selected as one of moduli m₀, m₁, . . . , m_(N-1). Ceiling[u] refers to a next highest whole integer with respect to an argument u.

In order to better understand the foregoing concepts, an example is useful. In this example, six (6) relatively prime moduli are used to solve six (6) irreducible polynomial equations f₀(x(nT)), . . . , f₅(x(nT)). A prime number p₀ associated with a first modulus m₀ is selected as five hundred three (503). A prime number p₁ associated with a second modulus m₁ is selected as four hundred ninety one (491). A prime number p₂ associated with a third modulus m₂ is selected as four hundred seventy-nine (479). A prime number p₃ associated with a fourth modulus m₃ is selected as four hundred sixty-seven (467). A prime number p₄ associated with a fifth modulus m₄ is selected as two hundred fifty-seven (257). A prime number p₅ associated with a sixth modulus m₅ is selected as two hundred fifty-one (251). Possible solutions for f₀(x(nT)) are in the range of zero (0) and five hundred two (502) which can be represented in nine (9) binary digits. Possible solutions for f₁(x(nT)) are in the range of zero (0) and four hundred ninety (490) which can be represented in nine (9) binary digits. Possible solutions for f₂(x(nT)) are in the range of zero (0) and four hundred seventy eight (478) which can be represented in nine (9) binary digits. Possible solutions for f₃(x(nT)) are in the range of zero (0) and four hundred sixty six (466) which can be represented in nine (9) binary digits. Possible solutions for f₄(x(nT)) are in the range of zero (0) and two hundred fifty six (256) which can be represented in nine (9) binary digits. Possible solutions for f₅(x(nT)) are in the range of zero (0) and two hundred fifty (250) which can be represented in eight (8) binary digits. Arithmetic for calculating the recursive solutions for polynomial equations f₀(x(nT)), . . . , f₄(x(nT)) requires nine (9) bit modulo arithmetic operations. The arithmetic for calculating the recursive solutions for polynomial equation f₅(x(nT)) requires eight (8) bit modulo arithmetic operations. In aggregate, the recursive results f₀(x(nT)), . . . , f₅(x(nT)) represent values in the range from zero (0) to M−1. The value of M is calculated as follows: p₀·p₁·p₂·p₃·p₄·p₅=503·491·479·467·257·251=3,563,762,191,059,523. The binary number system representation of each RNS solution can be computed using Ceiling[Log 2(3,563,762,191,059,523)]=Ceiling[51.66]=52 bits. Because each polynomial is irreducible, all 3,563,762,191,059,523 possible values are computed resulting in a sequence repetition time of every M times T seconds, i.e., a sequence repetition times an interval of time between exact replication of a sequence of generated values. Still, the invention is not limited in this regard.

Referring again to FIG. 9, the RNS solutions No. 1, . . . , No. N are mapped to a weighted number system representation thereby forming a chaotic sequence output. The phrase “weighted number system”, as used herein, refers to a number system other than a residue number system. Such weighted number systems include, but are not limited to, an integer number system, a binary number system, an octal number system, and a hexadecimal number system.

According to an aspect of the invention, the RNS solutions No. 1, . . . , No. N are mapped to a weighted number system representation by determining a series of digits in the weighted number system based on the RNS solutions No. 1, . . . , No. N. The term “digit”, as used herein, refers to a symbol of a combination of symbols to represent a number. For example, a digit can be a particular bit of a binary sequence. According to another aspect of the invention, the RNS solutions No. 1, . . . , No. N are mapped to a weighted number system representation by identifying a number in the weighted number system that is defined by the RNS solutions No. 1, . . . , No. N. According to yet another aspect of the invention, the RNS solutions No. 1, . . . , No. N are mapped to a weighted number system representation by identifying a truncated portion of a number in the weighted number system that is defined by the RNS solutions No. 1, . . . , No. N. The truncated portion can include any serially arranged set of digits of the number in the weighted number system. The truncated portion can also be exclusive of a most significant digit of the number in the weighted number system. The truncated portion can be a chaotic sequence with one or more digits removed from its beginning and/or ending. The truncated portion can also be a segment including a defined number of digits extracted from a chaotic sequence. The truncated portion can further be a result of a partial mapping of the RNS solutions No. 1, . . . , No. N to a weighted number system representation.

According to an embodiment of the invention, a mixed-radix conversion method is used for mapping RNS solutions No. 1, . . . , No. N to a weighted number system representation. “The mixed-radix conversion procedure to be described here can be implemented in” [modulo moduli only and not modulo the product of moduli.] See Residue Arithmetic and Its Applications To Computer Technology, written by Nicholas S. Szabo & Richard I. Tanaka, McGraw-Hill Book Co., New York, 1967. To be consistent with said reference, the following discussion of mixed radix conversion utilizes one (1) based variable indexing instead of zero (0) based indexing used elsewhere herein. In a mixed-radix number system, “a number x may be expressed in a mixed-radix form:

$x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; R_{i}}} + \ldots + {a_{3}R_{1}R_{2}} + {a_{2}R_{1}} + a_{1}}$

where the R_(i) are the radices, the a_(i) are the mixed-radix digits, and 0≦a_(i)<R_(i). For a given set of radices, the mixed-radix representation of x is denoted by (a_(n), a_(n-1), . . . , a₁) where the digits are listed in order of decreasing significance.” See Id. “The multipliers of the digits a_(i) are the mixed-radix weights where the weight of a_(i) is

${\prod\limits_{j = 1}^{i - 1}\; {R_{j}\mspace{14mu} {for}\mspace{14mu} i}} \neq {1.^{"}\mspace{14mu} {See}\mspace{14mu} {{Id}.}}$

For conversion from the RNS to a mixed-radix system, a set of moduli are chosen so that m_(i)=R_(i). A set of moduli are also chosen so that a mixed-radix system and a RNS are said to be associated. “In this case, the associated systems have the same range of values, that is

$\prod\limits_{i = 1}^{N}\; {m_{i}.}$

The mixed-radix conversion process described here may then be used to convert from the [RNS] to the mixed-radix system.” See Id.

“If m_(i)=R_(i), then the mixed-radix expression is of the form:

$x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}$

where a_(i) are the mixed-radix coefficients. The a_(i) are determined sequentially in the following manner, starting with a₁.” See Id.

$x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}$

is first taken modulo m₁. “Since all terms except the last are multiples of m₁, we have <x>_(m) ₁ =a₁. Hence, a₁ is just the first residue digit.” See Id.

“To obtain a₂, one first forms x−a₁ in its residue code. The quantity x−a₁ is obviously divisible by m₁. Furthermore, m₁ is relatively prime to all other moduli, by definition. Hence, the division remainder zero procedure [Division where the dividend is known to be an integer multiple of the divisor and the divisor is known to be relatively prime to M] can be used to find the residue digits of order 2 through N of

$\frac{x - a_{1}}{m_{1}}.$

Inspection of

$\left\lbrack {x = {{a_{N}{\prod\limits_{i = 1}^{N - 1}\; m_{i}}} + \ldots + {a_{3}m_{1}m_{2}} + {a_{2}m_{1}} + a_{1}}} \right\rbrack$

shows then that x is a₂. In this way, by successive subtracting and dividing in residue notation, all of the mixed-radix digits may be obtained.” See Id.

“It is interesting to note that

${a_{1}{\langle x\rangle}_{m_{1}}},{a_{2} = {\langle\left\lfloor \frac{x}{m_{1}} \right\rfloor\rangle}_{m2}},{a_{3} = {\langle\left\lfloor \frac{x}{m_{1}m_{2}} \right\rfloor\rangle}_{m_{3}}}$

and in general for i>1

${a_{i} = {\langle\left\lfloor \frac{x}{m_{1}m_{2}\mspace{14mu} \ldots \mspace{14mu} m_{i - 1}} \right\rfloor\rangle}_{m_{i}}}$

.” See Id. From the preceding description it is seen that the mixed-radix conversion process is iterative. The conversion can be modified to yield a truncated result. Still, the invention is not limited in this regard.

According to another embodiment of the invention, a Chinese remainder theorem (CRT) arithmetic operation is used to map the RNS solutions No. 1, . . . , No. N to a weighted number system representation. The CRT arithmetic operation can be defined by a mathematical equation (6) [returning to zero (0) based indexing].

$\begin{matrix} {{Y({nT})} = {\langle\begin{matrix} {{\left\lbrack {\langle{\left( {{3{x_{0}^{3}({nT})}3{x_{0}^{2}({nT})}} + {x_{0}({nT})} + C_{0}} \right)b_{0}}\rangle}_{p_{0}} \right\rbrack \frac{M}{p_{0}}} + \ldots +} \\ {\left\lbrack {\langle{\left( {{3{x_{N - 1}^{3}({nT})}} + {3{x_{N - 1}^{2}({nT})}} + {x_{N - 1}({nT})} + C_{N - 1}} \right)b_{N - 1}}\rangle}_{p_{N - 1}} \right\rbrack \frac{M}{p_{N - 1}}} \end{matrix}\rangle}_{M}} & (7) \end{matrix}$

where Y(nT) is the result of the CRT arithmetic operation; n is a sample time index value; T is a fixed constant having a value representing a time interval or increment; x₀−x_(N-1) are RNS solutions No. 1, . . . , No. N; p₀, p₁, . . . , p_(N-1) are prime numbers; M is a fixed constant defined by a product of the relatively prime numbers p₀, p₁, p_(N-1); and b₀, b₁, b_(N-1) are fixed constants that are chosen as the multiplicative inverses of the product of all other primes modulo p₀, p₁, . . . , p_(N-1), respectively.

Equivalently,

$b_{j} = {\left( \frac{M}{p_{j}} \right)^{- 1}{{{mod}p}_{j}.}}$

The b_(j)'s enable an isomorphic mapping between an RNS N-tuple value representing a weighted number and the weighted number. However without loss of chaotic properties, the mapping need only be unique and isomorphic. As such, a weighted number x can map into a tuple y. The tuple y can map into a weighted number z. The weighted number x is not equal to z as long as all tuples map into unique values for z in a range from zero (0) to M−1. Thus for certain embodiments of the present invention, the b_(j)'s can be defined as

$b_{j} = {\left( \frac{M}{p_{j}} \right)^{- 1}{{{mod}p}_{j}.}}$

In other embodiments of the present invention, all b_(j)'s can be set equal to one or more non-zero values without loss of the chaotic properties resulting in mathematical equation (7).

$\begin{matrix} {{Y({nT})} = {\langle\begin{matrix} {{\left\lbrack {\langle{{3{x_{0}^{3}({nT})}} + {3{x_{0}^{2}({nT})}} + {x_{0}({nT})} + C_{0}}\rangle}_{p_{0}} \right\rbrack \frac{M}{p_{0}}} + \ldots +} \\ {\left\lbrack {\langle{{3{x_{N - 1}^{3}({nT})}} + {3{x_{N - 1}^{2}({nT})}} + {x_{N - 1}({nT})} + C_{N - 1}}\rangle}_{p_{N - 1}} \right\rbrack \frac{M}{p_{N - 1}}} \end{matrix}\rangle}_{M}} & (7) \end{matrix}$

Embodiments of the present invention are not limited in this regard.

Referring again to FIG. 9, the chaotic sequence output can be expressed in a binary number system representation. As such, the chaotic sequence output can be represented as a binary sequence. Each bit of the binary sequence has a zero (0) value or a one (1) value. The chaotic sequence output can have a maximum bit length (MBL) defined by a mathematical equation (8).

MBL=Ceiling[Log 2(M)]  (8)

where M is the product of the relatively prime numbers p₀, p₁, . . . , p_(N-1) selected as moduli m₀, m₁, . . . , m_(N-1). In this regard, it should be appreciated the M represents a dynamic range of a CRT arithmetic operation. The phrase “dynamic range”, as used herein, refers to a maximum possible range of outcome values of a CRT arithmetic operation. It should also be appreciated that the CRT arithmetic operation generates a chaotic numerical sequence with a periodicity equal to the inverse of the dynamic range “M”. The dynamic range requires a Ceiling[Log 2(M)] bit precision.

According to an embodiment of the invention, M equals three quadrillion five hundred sixty-three trillion seven hundred sixty-two billion one hundred ninety-one million fifty-nine thousand five hundred twenty-three (3,563,762,191,059,523). By substituting the value of M into mathematical equation (8), the bit length (BL) for a chaotic sequence output expressed in a binary system representation can be calculated as follows: BL=Ceiling[Log 2(3,563,762,191,059,523)]=52 bits. As such, the chaotic sequence output is a fifty-two (52) bit binary sequence having an integer value between zero (0) and three quadrillion five hundred sixty-three trillion seven hundred sixty-two billion one hundred ninety-one million fifty-nine thousand five hundred twenty-two (3,563,762,191,059,522), inclusive. Still, the invention is not limited in this regard. For example, the chaotic sequence output can be a binary sequence representing a truncated portion of a value between zero (0) and M−1. In such a scenario, the chaotic sequence output can have a bit length less than Ceiling[Log 2(M)]. It should be noted that while truncation affects the dynamic range of the system it has no effect on the periodicity of a generated sequence.

As should be appreciated, the above-described chaotic sequence generation can be iteratively performed. In such a scenario, a feedback mechanism (e.g., a feedback loop) can be provided so that a variable “x” of a polynomial equation can be selectively defined as a solution computed in a previous iteration. Mathematical equation (4) can be rewritten in a general iterative form: f(x(nT)=Q(k)x³((n−1)T)+R(k)x²((n−1)T)+S(k)x((n−1)T)+C(k,L). For example, a fixed coefficient polynomial equation is selected as f(x(n·1 ms))=3x³((n−1)1 ms)+3x²((n−1)1 ms)+x((n−1)1 ms)+8 modulo 503. “n” is a variable having a value defined by an iteration being performed. “x” is a variable having a value allowable in a residue ring. In a first iteration, “n” equals one (1) and “x” is selected as two (2) which is allowable in a residue ring. By substituting the value of n and x into the stated polynomial equation f(x(nT)), a first solution having a value forty-six one (46) is obtained. In a second iteration, n is incremented by one and x equals the value of the first solution, i.e., forty-six (46) resulting in the solution 298, 410 mod 503 or one hundred thirty-one (131). In a third iteration, n is again incremented by one and x equals the value of the second solution.

Referring now to FIG. 10, there is provided a flow diagram of a method 1000 for generating a chaotic sequence according to an embodiment of the invention. It should be noted that the transmitters 102 ₁, . . . , 102 _(K) (described above in relation to FIG. 1-2 and FIG. 4) can be configured to generate different chaotic sequence (or orthogonal chaotic spreading codes CSC₁, . . . , CSC_(K)) using a chaotic sequence generation method 1000. As such, each of the transmitters 102 ₁, . . . , 102 _(K) can be provided with can be provided with different sets of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)), different sets of constants C₀, C₁, . . . , C_(N-1), and/or different sets of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected for use as modulus m₀, m₁, . . . , m_(N-1). The chaotic sequences generated at the transmitters 102 ₁, . . . , 102 _(K) provide orthogonal or statistically orthogonal chaotic spreading codes for spreading signals over a large common frequency band.

As shown in FIG. 10, the method 1000 begins with step 1002 and continues with step 1004. In step 1004, a plurality of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) are selected. The polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be selected as the same polynomial equation except for a different constant term or different polynomial equations. After step 1004, step 1006 is performed where a determination for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is made as to which combinations of RNS moduli m₀, m₁, . . . , m_(N-1) used for arithmetic operations and respective constant values C₀, C₁, . . . , C_(N-1) generate irreducible forms of each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). In step 1008, a modulus is selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) that is to be used for RNS arithmetic operations when solving the polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). The modulus is selected from the moduli identified in step 1006. It should also be appreciated that a different modulus must be selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)).

As shown in FIG. 10, method 1000 continues with a step 1010. In step 1010, a constant C_(m) is selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) for which a modulus is selected. Each constant C_(m) corresponds to the modulus selected for the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)). Each constant C_(m) is selected from among the possible constant values identified in step 1006 for generating an irreducible form of the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)).

After step 1010, method 1000 continues with step 1012. In step 1012, a value for time increment “T” is selected. Thereafter, an initial value for the variable “x” of the polynomial equations is selected. The initial value for “x” can be any value allowable in a residue ring. Notably, the initial value of “x” defines a sequence starting location. Subsequently, step 1016 is performed where RNS arithmetic operations are used to iteratively determine RNS solutions for each of the stated polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)). In step 1018, a series of digits in a weighted number system are determined based in the RNS solutions. Step 1018 can involve performing a mixed radix arithmetic operation or a CRT arithmetic operation using the RNS solutions to obtain a chaotic sequence output.

After step 1018, method 1000 continues with a decision step 1020. If a chaos generator is not terminated (1020:NO), then step 1024 is performed where a value of “x” in each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is set equal to the RNS solution computed for the respective polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) in step 1016. Subsequently, method 1000 returns to step 1016. If the chaos generator is terminated (1020:YES), then step 1022 is performed where method 1000 ends.

Referring now to FIG. 11, there is illustrated one embodiment of chaos generator 418. Chaos generator 418 is generally comprised of hardware and/or software configured to generate a digital chaotic sequence. Accordingly, chaos generator 418 is comprised of computing processors 1102 ₀, . . . , 1102 _(N-1) and a mapping processor 1104. Each computing processor 1102 ₀, . . . , 1102 _(N-1) is coupled to the mapping processor 1104 by a respective data bus 1106 ₀, . . . , 1106 _(N-1). As such, each computing processor 1102 ₀, . . . , 1102 _(N-1) is configured to communicate data to the mapping processor 1104 via a respective data bus 1106 ₀, . . . , 1106 _(N-1). The mapping processor 1104 can be coupled to an external device (not shown) via a data bus 1108. The external device (not shown) includes, but is not limited to, a communications device configured to combine or modify a signal in accordance with a chaotic sequence output.

Referring again to FIG. 11, the computing processors 1102 ₀, . . . , 1102 _(N-1) are comprised of hardware and/or software configured to solve N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) to obtain a plurality of solutions. The polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be irreducible polynomial equations having chaotic properties in Galois field arithmetic. Such irreducible polynomial equations include, but are not limited to, irreducible cubic polynomial equations and irreducible quadratic polynomial equations. The polynomial equations f₀(x(nT)) . . . f_(N-1)(x(nT)) can also be identical exclusive of a constant value. The constant value can be selected so that a polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is irreducible for a predefined modulus. The N polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)) can further be selected as a constant or varying function of time.

Each of the solutions can be expressed as a unique residue number system (RNS) N-tuple representation. In this regard, it should be appreciated that the computing processors 1102 ₀, . . . , 1102 _(N-1) employ modulo operations to calculate a respective solution for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) using modulo based arithmetic operations. Each of the computing processors 1102 ₀, . . . , 1102 _(N-1) is comprised of hardware and/or software configured to utilize a different relatively prime number p₀, p₁, . . . , p_(N-1) as a moduli m₀, m₁, . . . , m_(N-1) for modulo based arithmetic operations. The computing processors 1102 ₀, . . . , 1102 _(N-1) are also comprised of hardware and/or software configured to utilize modulus m₀, m₁, . . . , m_(N-1) selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) so that each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) is irreducible. The computing processors 1102 ₀, . . . , 1102 _(N-1) are further comprised of hardware and/or software configured to utilize moduli m₀, m₁, . . . , m_(N-1) selected for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) so that solutions iteratively computed via a feedback mechanism 1110 ₀, . . . , 1110 _(N-1) are chaotic. In this regard, it should be appreciated that the feedback mechanisms 1110 ₀, . . . , 1110 _(N-1) are provided so that the solutions for each polynomial equation f₀(x(nT)), . . . , f_(N-1)(x(nT)) can be iteratively computed. Accordingly, the feedback mechanisms 1110 ₀, . . . , 1110 _(N-1) are comprised of hardware and/or software configured to selectively define a variable “x” of a polynomial equation as a solution computed in a previous iteration.

Referring again to FIG. 11, the computing processors 1102 ₀, . . . , 1102 _(N-1) are further comprised of hardware and/or software configured to express each of the RNS residue values in a binary number system representation. In this regard, the computing processors 1102 ₀, . . . , 1102 _(N-1) can employ an RNS-to-binary conversion method. Such RNS-to-binary conversion methods are generally known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be appreciated that any such RNS-to-binary conversion method can be used without limitation. It should also be appreciated that the residue values expressed in binary number system representations are hereinafter referred to as moduli solutions No. 1, . . . , No. N comprising the elements of an RNS N-tuple.

According to an embodiment of the invention, the computing processors 1102 ₀, . . . , 1102 _(N-1) are further comprised of memory based tables (not shown) containing pre-computed residue values in a binary number system representation. The address space of each memory table is at least from zero (0) to m_(m)−1 for all m, m₀ through m_(N-1). The table address is used to initiate the chaotic sequence at the start of an iteration. The invention is not limited in this regard.

Referring again to FIG. 11, the mapping processor 1104 is comprised of hardware and/or software configured to map the moduli (RNS N-tuple) solutions No. 1, . . . , No. N to a weighted number system representation. The result is a series of digits in the weighted number system based on the moduli solutions No. 1, . . . , No. N. For example, the mapping processor 1104 can be comprised of hardware and/or software configured to determine the series of digits in the weighted number system based on the RNS residue values using a Chinese Remainder Theorem process. In this regard, it will be appreciated by those having ordinary skill in the art that the mapping processor 1104 is comprised of hardware and/or software configured to identify a number in the weighted number system that is defined by the moduli solutions No. 1, . . . , No. N.

According to an aspect of the invention, the mapping processor 1104 can be comprised of hardware and/or software configured to identify a truncated portion of a number in the weighted number system that is defined by the moduli solutions No. 1, . . . , No. N. For example, the mapping processor 1104 can be comprised of hardware and/or software configured to select the truncated portion to include any serially arranged set of digits of the number in the weighted number system. The mapping processor 1104 can also include hardware and/or software configured to select the truncated portion to be exclusive of a most significant digit when all possible weighted numbers represented by P bits are not mapped, i.e., when M−1<2^(P). P is a fewest number of bits required to achieve a binary representation of the weighted numbers. The invention is not limited in this regard.

Referring again to FIG. 11, the mapping processor 1104 is comprised of hardware and/or software configured to express a chaotic sequence in a binary number system representation. In this regard, it should be appreciated that the mapping processor 1004 can employ a weighted-to-binary conversion method. Weighted-to-binary conversion methods are generally known to persons having ordinary skill in the art, and therefore will not be described herein. However, it should be appreciated that any such weighted-to-binary conversion method can be used without limitation.

It should be noted that the transmitters 102 ₁, . . . , 102 _(K) are configured to generate different chaotic sequence (or orthogonal chaotic spreading codes CSC₁, . . . , CSC_(K)) using a chaotic sequence generation method as described above in relation to FIGS. 9-10. As such, each of the transmitters 102 ₁, . . . , 102 _(K) can be provided with can be provided with different sets of polynomial equations f₀(x(nT)), . . . , f_(N-1)(x(nT)), different sets of constants C₀, C₁, . . . , C_(N-1), and/or different sets of relatively prime numbers p₀, p₁, . . . , p_(N-1) selected for use as modulus m₀, m₁, . . . , m_(N-1). The chaotic sequences generated at the transmitters 102 ₁, . . . , 102 _(K) provide orthogonal or statistically orthogonal chaotic spreading codes for spreading signals over a large common frequency band. The spread spectrum signals can be transmitted from the transmitters to the base station 104 (described above in relation to FIG. 1) or receivers 154 ₁, . . . , 154 _(K) (described above in relation to FIG. 2). At the base station and/or receivers, the appropriate orthogonal spreading codes are used to recover the original signals intended for a particular user. Accordingly, the base station 104 is configured for generating replica chaotic sequences. The replica chaotic sequences are replicas of the orthogonal or statistically orthogonal chaotic spreading codes. Each of the replica chaotic sequences is synchronized in time and frequency with a respective one of the orthogonal chaotic spreading codes. Similarly, each of the receivers 154 ₁, . . . , 154 _(K) is configured to generate a replica chaotic sequence that is synchronized in time and frequency with a respective orthogonal chaotic spreading code.

All of the apparatus, methods, and algorithms disclosed and claimed herein can be made and executed without undue experimentation in light of the present disclosure. While the invention has been described in terms of preferred embodiments, it will be apparent to those of skill in the art that variations may be applied to the apparatus, methods and sequence of steps of the method without departing from the concept, spirit and scope of the invention. More specifically, it will be apparent that certain components may be added to, combined with, or substituted for the components described herein while the same or similar results would be achieved. All such similar substitutes and modifications apparent to those skilled in the art are deemed to be within the spirit, scope and concept of the invention as defined. 

1. A method for code-division multiplex communications, comprising: generating each of a plurality of orthogonal or statistically orthogonal chaotic spreading codes using a respective one of a plurality of different sets of polynomial equations; forming a plurality of spread spectrum communications signals respectively using said plurality of orthogonal or statistically orthogonal chaotic spreading codes; concurrently transmitting said plurality of spread spectrum communication signals over a common RF frequency band.
 2. The method according to claim 1, wherein said generating step further comprises using residue number system (RNS) arithmetic operations to respectively determine a plurality of solutions for each set of polynomial equations, said plurality of solutions iteratively computed and expressed as RNS residue values.
 3. The method according to claim 2, wherein said generating step further comprises determining a series of digits in a weighted number system based on said plurality of RNS residue values.
 4. The method according to claim 3, wherein said generating step further comprises selecting a value for each of N moduli comprising a respective moduli set in an RNS used for respectively solving each said set of polynomial equations.
 5. The method according to claim 4, wherein said moduli set is different for each said set of polynomial equations.
 6. The method according to claim 1, wherein said sets of polynomial equations differ with respect to at least one characteristic selected from the group comprising a constant and a polynomial degree.
 7. The method according to claim 1, wherein said generating step further comprises selecting a plurality of relatively prime numbers to be used as modulus in solving each set of polynomial equations.
 8. The method according to claim 1, further comprising: receiving said plurality of spread spectrum communications signals at a receiver; generating at least one chaotic de-spreading code; and de-spreading at least one of said spread spectrum communication signals using said chaotic de-spreading code.
 9. The method according to claim 1, further comprising: receiving said plurality of spread spectrum communications signals at a receiver; generating a plurality of chaotic de-spreading codes; and de-spreading a plurality of said spread spectrum communication signals using said plurality of chaotic de-spreading codes.
 10. The method according to claim 9, further comprising synchronizing said at least one of said plurality of chaotic de-spreading codes in time and frequency with at least one of said plurality of chaotic spreading codes.
 11. A method for code-division multiplex communications, comprising: generating a plurality of orthogonal or statistically orthogonal chaotic spreading codes using residue number system (RNS) arithmetic operations to respectively determine a plurality of solutions for each of a respective one of a plurality of sets of polynomial equations; selecting a different moduli sets for each set of polynomial equations, each moduli set comprising a value for each of N moduli in said RNS used for determining said plurality of solutions; forming a plurality of spread spectrum communications signals respectively using said plurality of orthogonal or statistically orthogonal chaotic spreading codes; and concurrently transmitting said plurality of spread spectrum communication signals over a common RF frequency band.
 12. The method according to claim 11, wherein said plurality of solutions are iteratively computed and expressed as RNS residue values.
 13. The method according to claim 12, wherein said generating step further comprises determining a series of digits in a weighted number system based on said plurality of RNS residue values.
 14. The method according to claim 11, wherein said sets of polynomial equations differ with respect to at least one characteristic selected from the group comprising a constant and a polynomial degree.
 15. The method according to claim 11, further comprising: receiving said plurality of spread spectrum communications signals at a receiver; generating at least one chaotic de-spreading code; and de-spreading at least one of said spread spectrum communication signals using said chaotic de-spreading code.
 16. The method according to claim 11, further comprising: receiving said plurality of spread spectrum communications signals at a receiver; generating a plurality of chaotic de-spreading codes; and de-spreading a plurality of said spread spectrum communication signals using said plurality of chaotic de-spreading codes.
 17. The method according to claim 16, further comprising synchronizing said at least one of said plurality of chaotic de-spreading codes in time and frequency with at least one of said plurality of chaotic spreading codes.
 18. A code-division multiplex communications system, comprising: a plurality of transmitters configured for (a) generating a plurality of orthogonal or statistically orthogonal chaotic spreading codes respectively using a plurality of different sets of polynomial equations, (b) forming a plurality of spread spectrum communications signals respectively using said plurality of orthogonal or statistically orthogonal chaotic spreading codes, and (c) concurrently transmitting said plurality of spread spectrum communication signals over a common RF frequency band.
 19. The code-division multiplex communications system according to claim 18, wherein each transmitter of said plurality of transmitters is further configured for generating a spreading code of said plurality of orthogonal or statistically orthogonal chaotic spreading codes using residue number system (RNS) arithmetic operations to respectively determine a plurality of solutions for a respective one of said plurality of different sets of polynomial equations, said plurality of solutions iteratively computed and expressed as RNS residue values.
 20. The code-division multiplex communications system according to claim 19, wherein each transmitter of said plurality of transmitters is further configured for generating a spreading code of said plurality of orthogonal or statistically orthogonal chaotic spreading codes by determining a series of digits in a weighted number system based on said plurality of RNS residue values.
 21. The code-division multiplex communications system according to claim 20, wherein each transmitter of said plurality of transmitters is further configured for generating a spreading code of said plurality of orthogonal or statistically orthogonal chaotic spreading codes by selecting a value for each of N moduli comprising a respective moduli set in an RNS used for respectively solving each said set of polynomial equations.
 22. The code-division multiplex communications system according to claim 21, wherein said moduli set is different for each said set of polynomial equations.
 23. The code-division multiplex communications system according to claim 18, wherein said sets of polynomial equations differ with respect to at least one characteristic selected from the group comprising a constant and a degree.
 24. The code-division multiplex communications system according to claim 18, wherein each transmitter of said plurality of transmitters is further configured for generating a spreading code of said plurality of orthogonal or statistically orthogonal chaotic spreading codes by selecting a plurality of relatively prime numbers to be used as modulus in solving each set of polynomial equations.
 25. The method according to claim 18, further comprising: receiving said plurality of spread spectrum communications signals at a receiver; generating a plurality of chaotic de-spreading codes; and de-spreading a plurality of said spread spectrum communication signals using said plurality of chaotic de-spreading codes; wherein said at least one of said plurality of chaotic de-spreading codes is synchronized in time and frequency with at least one of said plurality of chaotic spreading codes. 